This invention is related to carrier phase and symbol timing acquisition in time division multiple access (TDMA) burst communications and, or particularly, to the simultaneous acquisition of carrier phase and symbol timing synchronization from the preamble and and continued steady-state synchronization during the message burst of offset quaternary phase shift keying (O-QPSK) burst communications.
The modulation scheme usually employed in TDMA Satellite communications systems is conventional QPSK in which an in-phase carrier and quadrature carrier are each modulated with information and the modulated carriers are then combined into a single signal for transmission. A simplified block diagram of a QPSK signal generation technique is shown in FIG. 1 in which a carrier signal generator 10 provides one carrier signal to the input of multiplier 12 and also provides one input to the multiplier 14 through a 90 degree phase shift circuit 16. A timing signal generator 18 provides its output to a bit sequencer 20 which provides the polarity designations A and B to the mixers 12 and 14, respectively. Even and odd numbered information bits are conveyed by A and B, respectively, where a +1 corresponds to a logical "zero" and a 1 corresponds to a logical "one" for an information bit. The transition times for the polarity designations A and B are coincident.
Because only a single transmission is present at any time for a TDMA satellite transponder, it is feasible to operate the transponder at power saturation. However, problems arise with the use of QPSK signaling with non-linear satellite channels if the signal has significant spectral band limiting.
There are at least two basic conflicting features in TDMA satellite communications. First, the channel is non-linear and has a power constraint based upon peak rather than average capability. In order to take advantage of peak power capability, it is necessary for the signal to have an envelope that is nearly constant. However, the second feature of TDMA satellite communications is that the signal spectrum must be effectively truncated by filtering to avoid significant levels of out-of-band interference. For many modulation techniques, the intersymbol interference caused by filtering causes the envelope to have large variations. When the envelope variations are removed by either intentional amplitude limiting or by the limiting characteristic of the nonlinear channel, spectral side lobes are regrown that virtually remove the bandwidth constraint that was imposed by filtering. Thus, spectral band-limiting and constant-envelope signaling are incompatible for most modulation techniques when the satellite transponders and/or transmitter power amplifiers are operated in the non-linear regions near power saturation.
A necessary constaint on a transmission in order for it to accommodate both spectral band limiting and constant envelope is that band-limiting be accomplished by avoiding rapid phase transitions. In practical terms, this constraint must be imposed by a prohibition of polarity reversals or phase shifts of .pi. radian values. Such a .pi. phase shift occurs for conventional QPSK signals when both binary modulation components undergo transitions.
Due to this drawback in the use of QPSK signaling, an alternative modulation scheme referred to as "offset QPSK" (O-QPSK) is sometimes employed. In O-QPSK the timing of the binary sequences used to modulate the quadrature carrier components is staggered so that only one component at any one time may have a transition. Consequently, the phase shifts for O-QPSK are restricted to .+-..pi./2 radians. Filtering still results in considerable envelope variation of an O-QPSK transmission, but most of the spectral band limiting is manifested as a gradual rather than instantaneous phase shift when a bit transition occurs. Therefore, an O-QPSK signal can retain most of the spectral constraints imposed by filtering even if the envelope is restored to a constant value by either intentional means or by the saturated transponder response.
Although certain problems are solved by the use of O-QPSK, additional problems have been encountered in obtaining carrier phase and symbol timing synchronizations. The typical format for conventional TDMA/QPSK is to use a preamble to each burst that has alternating bit values for both binary modulation sequences A and B. Such a preamble allows the same overhead to be utilized for the acquisitions of both carrier phase and symbol timing. Usually, the carrier synchronization is obtained by double-squaring (4th power non-linearity) to remove the QPSK modulation, followed by a tuned filter at the fourth harmonic of the carrier and a divide-by-four zero-crossing counter. Symbol timing is obtained in parallel with carrier phase being acquired by a separate operation that employs a delay-and-multiply scheme and a filter tuned to the QPSK symbol rate R.sub.s. The delay is of one-half symbol duration, or 0.5/R.sub.s.
FIG. 2 illustrates a typical TDMA/QPSK synchronization scheme. The received QPSK signal is supplied to a filter 30 which is tuned to the carrier frequency f.sub.c, and the output of the filter 30 is supplied, in order, through a double squaring circuit 32, a filter 34 tuned to 4f.sub.c and a divide-by-four circuit 36. The output of the divide-by-four is supplied to a mixer 38 as the in-phase carrier for demodulation of the A-channel information, and it is also supplied through a 90 degree phase shifter 40 to the mixer 42 as the quadrature carrier for demodulation of the B-channel information. The outputs of mixers 38 and 42 are provided to respective lowpass filters 44 and 46 which are used for noise reduction, and the outputs of these filters are provided to sample-and-hold circuits 48 and 50. The sample and hold circuits are clocked by timing pulses synchronized to the symbol rate, and threshold detectors 52 and 54 provide the decoded bit decisions A and B.
In order to acquire symbol timing synchronization, the output of filter 30 is also provided as inputs to a mixer 56 and one-half symbol delay 58, the output of the delay 58 providing the second input to the mixer 56. The output of mixer 56 is provided to a filter 60 tuned to the QPSK symbol rate R and then to a timing pulse generator 62 which generates symbol rate timing pulses used to clock the sample-and-hold circuits 48 and 50.
For conventional QPSK, the two modulation sequences A and B have timing coincidence, and the transmission is defined by: ##EQU1## where C is the carrier power. During the preamble, the bit sequences A and B are coincident (A=B) and both sequences are alternating between +1 and -1 values. Consequently, the phase of the carrier will alternate between +45 degrees (.pi./4) and -135 degrees (-3.pi./4) during the preamble. Thus, the preamble is a form of binary phase shift keying (BPSK). Double-squaring of the QPSK signal and multiplication by -1 will result in all possible QPSK phase angles of .+-.45 degrees and .+-.135 degrees being rotated to 0 degrees, thereby producing an unmodulated signal at the fourth harmonic of the carrier. For the biphase preamble, however, only a single squaring in addition to a phase shift of -90 degrees will be required to remove the modulation, since the phase takes on only two values during that interval. Therefore, it is possible to acquire a carrier phase reference without the use of a fourth-power operation.
After squaring is used to acquire carrier phase during the BPSK preamble, a carrier reference will be available for coherent demodulation. Further, symbol timing will have been acquired, and bit decisions can be made. Note that if the carrier synchronizer of FIG. 2 is modified to include only a simple squaring circuit plus a -90 degree phase shift as discussed above, a coherent demodulated carrier can be obtained from the BPSK preamble, but the QPSK signal following the preamble will not be completely demodulated. Decision feedback (DFB) of the bit decisions can be used to remove the residual BSPK modulation in the signal at the second harmonic of the carrier that is obtained from squaring the QPSK transmission, so that squaring plus DFB allows carrier phase to be tracked throughout the remainder of the QPSK burst. TDMA system timing can be used to control the time at which DFB is employed, and the accuracy of switching to DFB is within a few symbol intervals, or a threshold detection on the output level of the carrier synchronizer filter can be used to control the DFB switch. Alternatively, the unique word at the end of the preamble may be designed such that its modulation scheme is only capable of occupying two states. In this way, the DFB need not be started until after the unique word, and a unique word detection signal could be used to trigger the DFB switch.
FIG. 3 is a block diagram of a typical TDMA/QPSK synchronization technique that employs squaring plus DFB for modulation removal, with similar components being designated by the same reference numerals as in FIG. 2. Instead of the fourth power circuit 32, 4f.sub.c filter 34 and divide-by-four circuit 36 in FIG. 2, the synchronization system of FIG. 3 squares the output of filter 30 in a mixer 70 and passes the output of mixer 70 through a filter 72 tuned to twice the carrier frequency. The output of filter 72 is provided through a delay adjustment 74 and phase adjustment 76 to one input of a mixer 78. During the preamble, the second input to mixer 78 is merely a +1 signal from the terminal P of switch 80, and the output of mixer 78 is provided through a filter 82 tuned to twice the carrier frequency, to a divide-by-two circuit 84 to obtain the coherent reference signal and, finally, to the mixer 38 and 90 degree phase shifter 40.
During the biphase preamble, squaring alone is employed for modulation removal, and after the preamble, squaring and DFB are employed with DFB being implemented in a simple re-modulation scheme. Squaring allows the carrier to be acquired at the second harmonic rather than at the fourth harmonic as is required for a fourth-power operation in FIG. 2 and, with low signal-to-noise E.sub.b /N.sub.o, squaring for modulation removal allows carrier phase synchronization to be obtained with much shorter preambles than when a fourth-power operation is employed. Further, there is a phase ambiguity of M states when an Mth-order non-linearity is used for modulation removal, and the ambiguity and carrier phase synchronization must be removed by detection of the phase state of a complex synchronization word or avoided by differential coding or other means. By creating only a two-state ambiguity, the use of squaring rather than fourth-power operation simplifies the ambiguity resolution.
The acquisition of symbol timing synchronization will now be described. If a and b denote the half-symbol delayed versions of the binary modulation sequences A and B, respectively, the two inputs to the mixer 56 are: EQU V.sub.1 (t)=A cos (w.sub.c t+.theta..sub.c)+B sin (w.sub.c t+.theta..sub.c) EQU V.sub.2 (t)=a cos (w.sub.c t+.theta..sub.c)+b sin (w.sub.c t+.theta..sub.c)
hence, the multiplier output is: ##EQU2##
Both aA and bB yield signals that can be filtered to obtain a timing signal, and since the A and B sequences have timing coincidence, the sum (aA+bB) is constructive. Accordingly, the lower side band of the product V.sub.1 V.sub.2 is passed through a filter 60 that is tuned to the symbol rate R.sub.s and is used to trigger timing pulses.
Although the above-described synchronization schemes work well for conventional QPSK, they cannot be used for O-QPSK synchronization. Due to the staggered timing of the bit sequences A and B, the products aA and bB will add destructively. Thus, although (aA+bB) is suitable for obtaining symbol timing for conventional QPSK, it is not at all suitable for symbol timing synchronization in O-QPSK. For O-QPSK, the desired signal for symbol synchronization would be (aA-bB). Note that (aA-bB) is the magnitude function for the cosine waveform at the second harmonic. Consequently, the correct signal for symbol synchronization on an O-QPSK transmission is superposed onto the carrier waveform. This interweaving of symbol timing and carrier phase into the same waveform is a characteristic of staggered QPSK, and one approach to synchronization for offset QPSK is to recognize the dependence of the carrier and symbol synchronization functions and not attempt to do independent synchronzations of each function. However, this requires that the carrier phase and symbol timing synchronization be acquired serially, and a longer preamble will be required. Rhodes et al "Computer Simulation of a Digital Satellite Communications System Utilizing TDMA and Coherent Quadriphase Signalling", Proceedings of ICC, 1972, pages 34-19 to 34-24, disclose a synchronization technique for O-QPSK in which carrier phase and symbol timing are updated during each TDMA burst rather than reacquired. Such a technique, however, will not be acceptable in reacquiring carrier phase and symbol timing for each burst, since the use of this technique would require serial acquisition of carrier phase and symbol timing. This technique is wasteful of overhead, with an unmodulated portion of the preamble required to acquire carrier phase before symbol timing can be acquired on a modulated portion of the preamble.
Rhodes, "Carrier Synchronization Techniques for Offset-QPSK Signals", National Telecommunications Conference Record San Diego, Dec. 2-4, 1974, pages 937-945 and Simon et al, "Offset Quadrature Communications With Decision Feedback Carrier Synchronization", IEEE Transactions on Communications, Vol. COM-22, No. 10, October 1974, pages 1576-1584, disclose synchronization techniques designed for O-QPSK signalling with continuous transmissions. However, synchronization circuits for continuous O-QPSK transmissions are not directly applicable to burst communications. In general, synchronization circuits for continuous transmissions have slow acquisition characteristics that would be inefficient for burst communications because of long preamble requirements.